Methods and Apparatus for Swept-Source Optical Coherence Tomography

ABSTRACT

In one embodiment of the invention, a semiconductor optical amplifier (SOA) in a laser ring is chosen to provide low polarization-dependent gain (PDG) and a booster semiconductor optical amplifier, outside of the ring, is chosen to provide high polarization-dependent gain. The use of a semiconductor optical amplifier with low polarization-dependent gain nearly eliminates variations in the polarization state of the light at the output of the laser, but does not eliminate the intra-sweep variations in the polarization state at the output of the laser, which can degrade the performance of the SS-OCT system.

RELATED APPLICATIONS

This application claims priority to U.S. Provisional Patent Application60/879,880 filed on Jan. 10, 2007, the disclosure of which is hereinincorporated by reference in its entirety.

FIELD OF INVENTION

This invention relates to the field of optical imaging and morespecifically to the design and implementation of optical coherencetomography (OCT) systems that employ swept-frequency lasers as lightsources.

BACKGROUND

Optical coherence tomography (OCT) is an interferometric imagingtechnique with widespread applications in ophthalmology, cardiology,gastroenterology and other fields of medicine. Huang D, Swanson E A, LinC P, Schuman J S, Stinson W G, Chang W, Hee M R, Flotte T, Gregory K,Puliafito C A, and Fujimoto J G, “Optical coherence tomography,”Science, Vol 254, 1178-1181 (1991). The ability to view subsurfacestructures with high resolution (2-15 μm) through small-diameterfiber-optic probes makes OCT especially useful for minimally invasiveimaging of internal tissues and organs. Commercially availabletime-domain OCT systems do not provide sufficient scan speed forunimpeded real-time visualization of organs that move rapidly or thathave large surface areas. In the beating heart, for example, OCT imagingof the coronary arteries is a challenge, because imaging must beperformed rapidly enough to allow clear visualization of a long segment(>3 cm) of an artery within the interval during which blood is clearedfrom the field of the view of the probe. The image acquisition rate ofthe current generation of commercially available OCT systems forcoronary artery imaging is limited to approximately 15 images/sec. Atthis acquisition speed, occlusion of the blood flow with a balloon forat least 30 seconds is required to image a 3-cm segment of the targetartery. If the image acquisition rate of OCT systems could be increasedby at least an order of magnitude, without significant loss of imagequality, balloon occlusion of long periods could be avoided. A segmentof an artery could then be imaged by simply injecting a bolus of salineover a few seconds, thereby simplifying the imaging procedure whilereducing the risk of myocardial ischemia.

Time-domain OCT systems employ a broadband light source as an input toan interferometer with a mechanically actuated reference arm forpath-length scanning. The interference signals generated by reflectionsfrom structures at different depths are measured point-by-point as thereference path length changes. In this measurement scheme, the maximumscanning speed is limited both by the dynamic mechanical constraints ofthe actuator and by the spectral power density of the light source. Insuch a system using a superluminescent light source that emits an outputpower of 25 mW over a spectral bandwidth of 40-60 nm, the maximumdepth-scanning velocity that can be achieved while maintaining anadequate signal-to-noise ratio for tissue imaging (>90 dB) isapproximately 25 m/s. Therefore, 512-line images of a 5 mm deep objectcan be acquired at a rate no greater than 10 per second.

Frequency-domain (also called Fourier-domain) (FD) OCT overcomes thesespeed constraints by taking advantage of optical frequencydiscrimination methods based on Fourier transformation, which eliminatethe need for long-range mechanical actuators. Swanson E A and Chinn S R,“Method and Apparatus for Performing Optical Frequency DomainReflectometry” U.S. Pat. No. 6,160,826 (issued Dec. 12, 2000); Choma MA, Sarunic M V, Yang C, and Izatt J, “Sensitivity advantage of sweptsource and Fourier domain optical coherence tomography,” Opt. Express,Vol. 11, 2183-2189 (2003). Instead of wasting available source power byinterrogating the sample point-by-point, FD-OCT collects informationfrom multiple depths simultaneously and discriminates reflections fromdifferent depths according to the optical frequencies of the signalsthey generate. FD-OCT imaging can be accomplished by illuminating thesample with a broadband source and dispersing the reflected light with aspectrometer onto an array detector. Alternatively, the sample can beilluminated with a rapid wavelength-tuned laser and the light reflectedduring a wavelength sweep collected with a single photodetector. In bothcases, a profile of reflections from different depths is obtained byFourier transformation of the recorded interference signals. Because oftheir potential to achieve higher performance at lower cost in the 1300nm spectral region, FD-OCT systems based on swept-frequency lasersources have attracted the most attention for medical applications thatrequire subsurface imaging in highly scattering tissues.

The feasibility of swept-source OCT (SS-OCT) has been demonstrated inseveral academic research studies. Chinn S R, Swanson E A, and FujimotoJ G, “Optical coherence tomography using a frequency-tunable opticalsource,” Opt. Lett., Vol. 22, 340-342 (1997); Yun S H, Tearney G J,Bouma B E, Park B H, de Boer J F, “High-speed spectral domain opticalcoherence tomography at 1.3 μm wavelength,” Optics Express, Vol. 11, pp.3598-3604 (2003); Choma M A, Hsu K, and Izatt J, “Swept source opticalcoherence tomography using an all-fiber 1300 nm ring laser source,” J.Biomed. Optics, Vol. 10, p. 044009 (2005); Huber R, Wojtkowski, Taira K,Fujimoto J G, and Hsu K, “Amplified, frequency-swept lasers forfrequency domain reflectometry and OCT imaging: design and scalingprinciples,” Opt. Express, Vol. 13, 3513-3528 (2005). Most of thereported SS-OCT systems employ short-cavity lasers tuned rapidly by anelectronically actuated Fabry-Perot filter or a motor-driven gratingfilter. The implementations disclosed to date suffer from drawbacks thatdiscourage widespread commercialization of SS-OCT. Specifically, currentimplementations make real-time data acquisition and display difficult,because they employ data acquisition schemes that requirepost-acquisition re-sampling or interpolation of recorded data beforeFourier transformation. In addition, the relatively short coherencelength and tendency for mode-hopping of short-cavity lasers reducesignal-to-noise and image resolution at optical scan depths exceeding2-3 mm. Many medical applications, including coronary artery imaging,require an optical scan depth that exceeds 5 mm.

The recent development of Fourier-Domain Mode Locking (FDML) solves theproblem of degraded signal-to-noise and image resolution at largeoptical scan depths. Huber R, Taira K, and Fujimoto J, “Mode LockingMethods and Apparatus,” U.S. Patent Application No. 2006/0187537,(Published Aug. 24, 2006); Huber R, Wojtkowski M, and Fujimoro J G,“Fourier Domain Mode Locking (FDML): A new laser operating regime andapplications for optical coherence tomography,” Optics Express, Vol. 14,pp. 3225-3237 (2006). However, the practical implementation of aFDML-based SS-OCT system presents several technical challenges. Thepresent invention addresses these challenges and provides solutions tothe same.

SUMMARY OF THE INVENTION

The present invention describes devices and methods that enable stable,low-noise, and efficient operation of swept-source OCT (SS-OCT) systemsat high speed, with continuous real-time image display. The methodsdetailed herein overcome disadvantages of previous implementations ofSS-OCT systems, which include poor noise performance, limited scanrange, the effects the birefringence and dispersion properties of thelaser cavity, phase jitter, and sampling speed limitations.

In one aspect, the invention relates to an optical coherence tomographydata collection apparatus. The apparatus can include a first gainelement, a second gain element, where each gain element has a differentgain dependence on polarization, and a Fourier-domain mode-locked laserdefining a cavity. The laser can include a frequency tuning element inoptical communication with the first gain element, where the first gainelement can be disposed within the laser cavity and the second gainelement can be disposed outside the cavity, and the gain dependence onpolarization of the first gain element is less than the gain dependenceon polarization of the second gain element.

The apparatus can include a sample clock generator, where the sampleclock generator can be configured to clock an analog-to-digitalconverter. The analog-to-digital converter can be configured to sampleinterference signals at an output of a main interferometer. Theapparatus can include a digital control system configured to stabilize adrive frequency of the frequency tuning element of the Fourier-domainmode-locked laser using at least one control signal derived from thesample clock generator. The Fourier-domain mode-locked laser can includean optical delay element that includes a pair of fiber coils whoserelative orientations are adjusted to reduce the effects of polarizationmode dispersion.

The sample clock generator can include a sample clock interferometer, aphotoreceiver, an automatic gain control amplifier, a frequencymultiplier, a zero-crossing detector, and/or a clock switch. The sampleclock generator can include a Mach-Zehnder interferometer, including apair of 2×2 fiber couplers, a Michelson interferometer with mismatchedlengths in sample and reference arms, a common-path Michelsoninterferometer, including an element with two partially reflectiveinterfaces, and/or a Fabry-Perot interferometer. The sample clockgenerator can include an analog multiplier. The analog multiplier can beconfigured to perform a squaring function on an input interferencesignal. The sample clock generator can include an analog multiplier forthe multiplication of a pair of signals derived from an interferencesignal transmitted through a phase-shifting RF power splitter. Thesample clock generator can include an exclusive OR gate for thetransmission of a pair of phase-shifted pulse trains, the pulse trainsderived from a zero-crossing detector applied to an interference signal,and a delayed replica of the zero-crossing detector's output. The sampleclock generator can include an exclusive OR gate for the transmission ofa pair of phase-shifted pulse trains, wherein the pulse trains arederived from a pair of zero-crossing detectors applied to sinusoidalsignals derived from a phase-shifting power splitter.

The sample clock interferometer can generate phase-shifted interferencesignals for frequency modulation from a combination of a 2×2 coupler anda 3×3 coupler. The power splitting ratio of the 3×3 coupler can bechosen to obtain a pair of interference signals whose phases differ byapproximately 90 degrees. The sample clock interferometer can generatephase-shifted interference signals for frequency modulation from acombination of a 2×2 coupler and a 3×3 coupler, the power splittingratio of the 3×3 coupler chosen to obtain a pair of interference signalswhose phases differ by approximately 90 degrees. The apparatus canfurther include a 4×4 coupler, the 4×4 coupler generating a pair ofbalanced signals with a quadrature phase relationship, the sample clockgenerator generating a single ADC clock signal. The sample clockgenerator can generate complex-valued signals for Fourier transformationby recording OCT data using a pair of ADC clock signals whose phasesdiffer by 90 degrees.

In one aspect, the invention relates to a method of OCT imaging. Themethod can include generating light from a Fourier-domain mode-lockedlaser, where the laser can define a cavity and include a first gainelement, and the first gain element can be disposed within the lasercavity. The method can include transmitting the generated light througha second gain element, where the second gain element can be disposedoutside the cavity and each gain element can have a different gaindependence on polarization. The gain dependence on polarization of thefirst gain element can be less than the gain dependence on polarizationof the second gain element. The method can include sampling interferencesignals at an output of a main interferometer using an analog-to-digitalconverter. The analog-to-digital converter can be clocked using a sampleclock generator. The method can include optimizing a drive frequency ofa frequency tuning element of the laser with a digital control system,where at least one control signal derived from the sample clockgenerator.

The method can further include the step of generating a pair of balancedsignals with a quadrature phase relationship for dual-channelacquisition of OCT signals from the main interferometer using a 4×4coupler, whereby only a single ADC clock signal from the sample clockgenerator is required. The step of optimizing the drive frequency caninclude measuring the instantaneous RMS amplitude Φ(t) of theinterference signal at the output of the sample clock interferometer'sphotoreceiver at the time τ indicated by transmission of a pulse througha fiber-Bragg filter with a narrow passband in the vicinity of thezero-dispersion wavelength of the optical delay element and adjustingthe frequency of a direct digital synthesis (DDS) generator to maximizethe value of Φ(t).

The step of optimizing the drive frequency can include measuring thedesired delay D between zero-crossing times of a drive waveform and aninitial laser sweep and adjusting a dc bias with a digital-to-analogconverter to maintain a fixed delay τ−D, where τ is the time measured bytransmission of the pulse through a fiber-Bragg filter with a narrowpassband in the vicinity of the zero-dispersion wavelength of theoptical delay element.

In another aspect, the invention relates to an optical coherencetomography data collection apparatus, the apparatus includes aninterferometer having an input and an output, an analog-to-digitalconverter configured to sample interference signals from the output, afirst gain element, a second gain element, where each gain element canhave a different gain dependence on polarization, a Fourier-domainmode-locked laser having a laser cavity, where the laser can be inoptical communication with the interferometer. The laser can include afrequency tuning element in optical communication with the first gainelement, the first gain element disposed within the laser cavity, thesecond gain element disposed outside the cavity, a sample clockgenerator configured to clock an analog-to-digital converter, and adigital control system configured to stabilize a drive frequency of thefrequency tuning element of the laser, using a control signal derivedfrom the sample clock generator. The gain dependence on polarization ofthe first gain element can be less than the in dependence onpolarization of the second gain element. The laser can include anoptical delay element that includes a pair of fiber coils whose relativeorientations are adjusted to reduce the effects of polarization modedispersion.

The sample clock generator can include a sample clock interferometer, aphotoreceiver, an automatic gain control amplifier, an optionalfrequency multiplier, a zero-crossing detector, and/or a clock switch.The sample clock generator can include an analog multiplier, where theanalog multiplier can be configured to perform a squaring function on aninput interference signal.

In one aspect, the invention relates to a method of increasing a usefulduty cycle of a tuning element in a cavity of a FDML laser. The methodincludes the steps of linearizing a portion of a frequency tuningelement duty cycle and driving a filter. The filter combines a pluralityof phase-locked sine waves having a harmonic frequency relation, eachwave having an adjustable amplitude and phase. In one embodiment, two ofthe plurality of sine waves having frequencies f and 2f are generated bya pair of phase-locked digital-direct synthesis integrated circuits andwhose weighted sum yields a smoothed ramp-like displacement of apiezo-electric or micro-electromechanical systems (MEMs) Fabry-Perottunable filter. In another embodiment, two of the plurality of sinewaves having frequencies f and 3f are generated by a pair ofphase-locked digital-direct synthesis integrated circuits and whoseweighted sum yields a triangular-wave displacement of a piezo-electricor micro-electromechanical systems (MEMs) Fabry-Perot tunable filter.

In one embodiment of the invention, a semiconductor optical amplifier(SOA) in a laser ring is chosen to provide low polarization-dependentgain (PDG) and a booster semiconductor optical amplifier, outside of thering, is chosen to provide high polarization-dependent gain. The use ofa semiconductor optical amplifier with low polarization-dependent gainnearly eliminates variations in the polarization state of the light atthe output of the laser, but does not eliminate the intra-sweepvariations in the polarization state at the output of the laser, whichcan degrade the performance of the SS-OCT system. Some of theembodiments disclosed herein overcome variations in both the amplitudeand polarization, because the booster semiconductor optical amplifierprovides sufficient amplification in a single polarization axis to reachgain saturation at all wavelengths, in spite of the polarizationvariations at the output of the low polarization-dependent gain ringsemiconductor optical amplifier.

One objective of this invention is to present methods for stabilizationof the polarization state of light circulating within the long-path ringcavity of an FDML laser. These methods improve the performance andmanufacturability of SS-OCT systems based on the FDML principle.

Another objective of the present invention is to describeopto-electronic methods and devices for generating a stable sample clockfor direct acquisition of interferometric systems from various types ofSS-OCT systems. These methods reduce phase noise, expand the dynamicrange, and increase the acquisition speed of the acquired interferencesignals.

A further objective of the present invention is to disclose methods anddevices for feedback stabilization of FDML SS-OCT systems. A practicalembodiment is presented that includes a frequency-agile, direct digitalsynthesis (DDS) waveform generator and a digital microcontrollerconfigured for optimization of an electronic feedback variable. Acompanion method for stabilization of the starting wavelength of thewavelength sweep of the FDML laser is also disclosed.

A still further objective of the present invention is to disclosemethods for linearizing and extending the duty cycle of the opticalfrequency sweep of tunable lasers. These methods, which operate at highscan repetition rates, can be applied to piezoelectric andmicroelectromechanical (MEMs) actuators, including, but not limited toactuators with highly resonant mechanical properties.

Another further objective of the present invention is to reduce foldoverartifacts. In one embodiment, a foldover artifact occurs when a sampleportion under investigation is projected upon an opposing side of thesample portion to result in ambiguities in any resultant image. As usedherein, a foldover artifact also refers to any phase wrapping, wraparound, or aliasing based ambiguities associated with OCT data capture.

The methods and systems are explained through the following description,drawings, and claims.

BRIEF DESCRIPTION OF THE DRAWINGS

The objects and features of the invention can be understood morecompletely by referring to the drawings described below and theaccompanying descriptions. The drawings are not necessarily to scale,emphasis instead generally being placed upon illustrating the principlesof the invention. In the drawings, like numerals are used to indicatelike parts throughout the various views.

FIG. 1 is a block diagram of an SS-OCT system according to anillustrative embodiment of the invention.

FIG. 2 shows a preferred embodiment of a FDML laser, configured toproduce an output with high polarization stability according to anillustrative embodiment of the invention.

FIG. 3 shows an alternative design for the optical delay element in FIG.2 that includes a pair of optical fiber coils oriented at an angle α˜90degrees for reducing polarization mode dispersion according to anillustrative embodiment of the invention.

FIG. 4 shows a general embodiment of a sample clock generator accordingto an illustrative embodiment of the invention.

FIG. 5 shows two specific embodiments, 5(a) and 5(b), of the frequencymultiplier in FIG. 4 according to illustrative embodiments of theinvention.

FIG. 6 shows two additional embodiments, 6(a) and 6(b), of frequencymultipliers according to illustrative embodiments of the invention.

FIG. 7 is another embodiment of the sample clock generator in which theMach-Zehnder sample clock interferometer is replaced by a 3×3phase-splitting interferometer according to an illustrative embodimentof the invention.

FIG. 8 is a modified version of the sample clock generator shown in FIG.7, in which the two quadrature outputs are first passed thoughzero-crossing detectors and then exclusive-ORed to generated thefrequency-doubled ADC clock according to an illustrative embodiment ofthe invention.

FIG. 9 is yet another embodiment of the sample clock generator accordingto an illustrative embodiment of the invention.

FIG. 10 shows a configuration in which the quadrature sample clocks areemployed as separate clocks for quadrature detection of OCT signals fromthe main interferometer according to an illustrative embodiment of theinvention.

FIG. 11 depicts the application of a 4×4 optical coupler fordual-channel acquisition of balanced quadrature OCT signals from themain interferometer according to an illustrative embodiment of theinvention.

FIG. 12 illustrates a specific embodiment of a digital feedback loop foroptimization and stabilization of the drive frequency of an FDML-basedSS-OCT system according to an illustrative embodiment of the invention.

FIG. 13 shows the measured amplitudes and shapes of the clock fringecontrol signal (RMS fringe amplitude) for optimum and non-optimum (toolow or too high) adjustment of the frequency of the waveform driving thefrequency tuning element in the FDML laser according to one specificembodiment of the invention.

FIG. 14 illustrates a specific embodiment of a feedback control loop foroptimization and stabilization of the dc bias voltage applied to thetuning element in the FDML laser according to an illustrative embodimentof the invention.

FIG. 15 shows the typical highly resonant frequency response of apiezo-actuated Fabry-Perot tunable filter according to an illustrativeembodiment of the invention.

FIG. 16 illustrates the principle of Fourier synthesis on which anexemplary actuator linearization method is based according to anillustrative embodiment of the invention.

FIG. 17 shows an example of a configuration for synthesized harmoniclinearization of a piezo-actuated Fabry-Perot tunable filter accordingto an illustrative embodiment of the invention.

FIG. 18 shows a specific embodiment of a tunable-filter linearizingcircuit based according to an illustrative embodiment of the invention.

DETAILED DESCRIPTION

The following description refers to the accompanying drawings thatillustrate certain embodiments of the invention. Other embodiments arepossible and modifications may be made to the embodiments withoutdeparting from the spirit and scope of the invention. Therefore, thefollowing detailed description is not meant to limit the invention.Rather, the scope of the invention is defined by the appended claims.

In general, the invention relates to apparatus and methods for enhancedswept-source OCT system suitable for imaging various structures, such asthose in a cadaver or living organism. Typically, these systems arebased on a Fourier-Domain Mode Locking (FDML) approach. UsingFourier-Domain Mode Locking (FDML) when implementing the systems andmethods described herein solves the problem of degraded signal-to-noiseratio and image resolution at large optical scan depths. However, thepractical implementation of an FDML-based SS-OCT system presents severaltechnical challenges.

First, to ensure stable and low-noise operation of an FDML-based SS-OCTsystem, the effects of the birefringence and dispersion properties ofthe laser cavity must be minimized. Second, to maintain thefrequency-mode-locked condition, the period of the waveform that drivesthe tunable filter must have extremely low-phase jitter and must bematched precisely to the round-trip delay through the laser cavity. Ifthe period of the drive waveform and round-trip delay differ by morethan a small fraction (e.g., 10 ppm), the coherence and noise propertiesof the laser degrade markedly.

Moreover, to compensate for environmental influences, the period of thedrive waveform must change in response to changes in the length of thecavity. Third, to ensure repeatable phase and amplitude characteristicsof the acquired interference signals, the wavelength from which thewavelength sweep starts must be kept the same from sweep to sweep.Fourth, to enable real-time operation a FDML laser, configured toproduce an output with high resolution, the interference signals must besampled at high speed at precise optical-frequency intervals.

Aspects of the invention describe devices and methods that address theproblems identified above by incorporating specific components in andadjusting the overall configuration of various FDML-based SS-OCTsystems. Accordingly, the methods and apparatus described herein enablestable, low-noise, and efficient operation of swept-source OCT systemsat high speed, with continuous real-time image display. The methodsdetailed herein overcome disadvantages of previous implementations ofSS-OCT systems, which include high system cost and complexity, poornoise performance, and limited scan range.

In particular, since one or more long optical fiber loops are used insome of the systems disclosed herein to match the travel time in anoptical circuit with the switching time of an electric circuit,environmental influences such as temperature variations and mechanicalstress can introduce unwanted polarization effects in the optical fiberloops. In part, the embodiments disclosed herein overcome variations inboth the amplitude and polarization, through implementation of boostersemiconductor optical amplifiers to provide sufficient amplification ina single polarization axis to reach gain saturation at all wavelengths,in spite of the polarization variations at the output of the lowpolarization-dependent gain ring semiconductor optical amplifier. Thus,the presence of the semiconductor optical amplifiers address theproblems introduced by providing long runs of optical fiber to matchoptical travel and electronic switching times.

General aspects of the invention and various embodiments illustratingsystems and methods that address the problems recited above aredescribed in more detail with respect to the accompanying figures. FIG.1 illustrates a general implementation of a FDML based system S₁suitable for use with an OCT probe. In turn, FIG. 2 provides specificdetails relating to the use of gain elements, such as, but not limitedto semiconductor optical amplifiers to counteract unwanted polarizationeffects. Further, the system of FIG. 3 shows an alternative design forthe optical delay element in FIG. 2 that includes a pair of opticalfiber coils oriented at an angle α˜90 degrees for reducing polarizationmode dispersion according to an illustrative embodiment of theinvention.

Returning to FIG. 1, a swept-source OCT (SS-OCT) system S₁ havingvarious specific components is depicted. The main components of thesystem on which an embodiment of the invention is based are shown inFIG. 1. Specifically, FIG. 1 includes a tunable (wavelength-swept) laserL that includes an optical delay element 1; an optical frequency tuningelement 2 a; and a first gain element 2 b The optical frequency tuningelement 2 a has one or more control inputs from a frequency-agile directdigital-synthesized waveform generator 3; and a digital-to-analogconverter 4. This system S₁ is configured to achieve FDML and providethe benefits discussed above. Light, from the laser L, travels to a maininterferometer which is in optical communication with an OCT probe.Light received from the OCT probe is transmitted back to the maininterferometer and captured by a photoreceiver and ultimately convertedto scan data.

As shown in FIG. 1, a microcontroller 5 for laser stabilization thatreceives a wavelength synchronization (λ sync) input from a fiber Bragggrating filter 6 is also part of the system S₁. A sample clock generator7 provides the sample clock directly to the main analog-to-digitalconverter (ADC). The system also includes a clock fringe control inputfrom the sample clock generator 7 that is in electrical communicationwith the microcontroller 5. In general, all of the elements showh inFIG. 1 are in electrical or optical communication along the paths shown,as appropriate for a given embodiment.

As shown in the figure, light from an FDML laser L is split into areference and sample path by the main interferometer. The electronicinterference signal is detected by a balanced photoreceiver. In turn,the photoreceiver's output signal is processed at high speed by the mainADC. A small fraction of the light from the laser L enters the sampleclock generator 7, which produces 1) a low-jitter sample clock for themain ADC and a 2) clock fringe signal that serves as the controlvariable for stabilization of the ac drive waveform of the frequencytuning element 2 a.

Examples of frequency tuning elements include piezo-actuated Fabry-Perotfilters and galvanometer-actuated grating filters. Another smallfraction of the light from the laser L passes through a narrowband fiberBragg grating filter into a third photoreceiver that generates awavelength sync pulse. This sync pulse serves as the reference timemarker for controlling the dc bias voltage of the frequency tuningelement. The microcontroller performs the data acquisition and digitalprocessing tasks associated with feedback control of the frequency of acdrive waveform and dc bias voltage. The ac drive frequency is controlledvia a digital control word (typically 4 bytes or more) generated by themicrocontroller to the direct digit synthesis (DDS) waveform synthesizer(e.g., Analog Devices AD9952).

Typically, the DDS synthesizer 3 is configured to generate a sinusoid inthe 20-100 KHz range, whose frequency can be altered rapidly with aresolution better than 0.05 Hz. To produce a waveform with extremely lowjitter, a high-frequency (typically>100 MHz), high stability (<10 ppm)oscillator, such as a crystal oscillator, can be used as the baselineclock for the DDS synthesizer 3. An additional digital control wordgenerated by the embedded microcontroller and transmitted to adigital-to-analog converter (4), controls the dc bias of the frequencytuning element.

In contrast to the general overall system of FIG. 1, FIG. 2 shows apreferred embodiment of an FDML laser, configured to provide an outputwith high polarization stability. The embodiment of FIG. 2 can be usedin the system of FIG. 1. The FDML laser of FIG. 2 addresses theproblematic polarization effects introduced by mechanical and thermalstresses discussed above. Although the general layout is similar to thatdescribed by Huber et al (U.S. Patent Application No. 2006/0187537), thefirst and second gain elements are chosen to satisfy specificrequirements. In particular, the semiconductor optical amplifier (SOA)in the optical fiber ring (cavity), the first gain element, is chosen toprovide low polarization-dependent gain (PDG). In turn, the boostersemiconductor optical amplifier, an exemplary second gain element, ischosen to provide high polarization-dependent gain. The use of the terms“low” and “high” with respect to polarization-dependent gain (PDG)elements indicate the relative level of polarization-gain dependencesuch the polarization dependence of the high PDG element is greater thanthe polarization dependence of the low PDG element.

In one embodiment, a gain element, such as an amplifier, with a PDG lessthan about 3 dB can be considered a low PDG gain element. Conversely, inone embodiment, a gain element, such as an amplifier, with a PDG greaterthan or equal to about 3 dB can be considered a high PDG gain element.Further, in this context, a 3 dB PDG means that the two orthogonalpolarization states are amplified to within 3 dB of each other.

In a conventional arrangement in which only a single SOA (either ahigh-PDG or low-PDG version) is used inside the ring or the SOAs withsimilar PDGs are used for both the ring or booster, large variations inthe light's polarization state at the laser's output occur as the lasersweeps across a wide band of wavelengths. The wavelength dependence ofthe polarization-mode dispersion (PMD) within the optical delay elementand the other optical elements inside the ring are the likely source ofthese effects. It is worth noting that the low PDG SOA does noteliminate the intra-sweep variations in the polarization state at theoutput of the laser, which can degrade the performance of the SS-OCTsystem.

The configuration described in FIG. 2 overcomes variations in both theamplitude and polarization, because the booster SOA (second gainelement) provides sufficient amplification in a single polarization axisto reach gain saturation at all wavelengths, in spite of thepolarization variations at the output of the low-PDG ring SOA (firstgain element).

Turning now to FIG. 3, an alternative design for the optical delayelement of FIG. 2 is shown. Specifically, in FIG. 3 the delay elementshown includes a pair of optical fiber coils oriented at an angle φ˜90degrees for reducing polarization mode dispersion. The split-coilarrangement of the fiber optic delay element shown in FIG. 3 is designedto further reduce the effects of PMD inside the optical fiber ring thatis used to match optical travel and electronic switching times in a FDMLsystem. By orienting the coils at an angle φ that is substantially equalto 90 degrees, the group-delay difference between the orthogonalpolarization modes in the first coil is compensated by an opposingdifference in the second coil. This compensating effect results from theorthogonal orientations of the birefringence axes of the two coils.Thus, the embodiment shown in FIG. 3 further reduces unwantedpolarization effects in the larger optical ring.

In general, aspects of the invention relate to the selection andmatching of components for use in an FDML OCT system. The selection ofthe sample clock generator is another aspect of the invention. As shownin FIG. 1, the sample clock generator 7 is in communication withdifferent controls and the FDML laser. The function of the sample clockis twofold. First, it is used to generate a sample clock for the mainanalog-to-digital converter and, second, to generate a clock fringecontrol signal for use by the microcontroller 5.

As shown in FIG. 1, the microcontroller 5 uses the clock fringe controlsignal to determine a substantially optimum drive frequency forcontrolling the frequency tuning element connected to or integratedwithin the FDML laser. The sample clock generator derives low-jitterclock pulses from the sinusoidal interference signals generated by thesample clock interferometer. Although the time intervals of the clockpulses vary as the wavelength of the laser sweeps, equal spacing of theintervals between the clock edges in the optical frequency domain ismaintained. These characteristics allow direct clocking of certain typesof high-speed analog-to-digital converters, such as flash A/D convertersor pipelined A/D converters that accept a variable frequency clock(e.g., AD9340), without the need for complex resampling hardware. Thus,given the significance of synchronizing optical trains in a FDML systemthe selection of the clock generator and various enhancements relatingthereto improve the overall quality of the scan data obtained from anOCT probe. Additional details relating to sample clock generatorembodiments are described and/or shown in more detail below with respectto FIGS. 4-10.

FIG. 4 shows a general embodiment of a sample clock generator 8 thatderives a stable analog-to-digital converter (ADC) clock from thebalanced outputs of a Mach-Zehnder interferometer 10. The frequencymultiplier (4) (M=2, 3, . . . ) permits ADC clocking at rates higherthan the fundamental frequency of the Mach-Zehnder interference signals.In one embodiment, the generator includes an optional set of componentssuch as a crystal oscillator 12 and RF clock switch 13 that permit theuse of analog-to-digital converters that provide non-interruptedclocking.

As depicted, FIG. 4 illustrates the basic configuration of the sampleclock generator 8. A photoreceiver converts the optical interferencesignals from the sample clock interferometer shown in this embodiment asthe Mach-Zehnder interferometer 10 with an optical path imbalance equalto ΔL, into a chirped sinusoidal waveform. The waveform is filtered topass the band of frequencies generated by sweeping the FDML laserbetween its wavelength limits. To equalize the amplitude of theinterference signals generated during the sweep and to reduce phaseerrors after zero-crossing detection, the filtered waveform passesthorough an amplifier with automatic gain control (AGC).

An optional frequency multiplier 14 multiplies the frequency of theband-passed waveform, typically by a factor of 2 to 4. The frequencymultiplier 14 (M=2, 3, . . . ) permits ADC clocking at rates higher thanthe fundamental frequency of the Mach-Zehnder interference signals.Because it allows swept-source lasers to generate synchronous ADCclocking rates above the Nyquist frequency when the path lengthimbalance is set equal to the coherence length of the laser, frequencymultiplication enhances the operation of clock generators designed foruse with high-resolution SS-OCT systems with long scan ranges. Afterfrequency multiplication, the waveform is filtered again to eliminateundesired harmonics and the residual signal components at thefundamental frequency.

In turn, in the embodiment of FIG. 4, a zero-crossing detector convertsthe waveform into a pulse train with variable spacing in the timedomain, but equal spacing in the optical frequency domain. An optionalclock switch, composed of a crystal oscillator and RF switch, interposesa fixed frequency pulse train between variable-frequency pulse trainsgenerated during the periodic sweep interval. The clock switch permitsthe use of analog-to-digital converters that require non-interruptedclocking.

Two alternative embodiments of the frequency multiplier of FIG. 4 aredepicted in FIGS. 5 a and 5 b. Specifically, the two frequencymultiplier embodiments shown are designed for doubling (M=2) thefrequency of sinusoidal interference signals with frequencies that sweepover the range f_(L) to f_(H) during the acquisition period according toillustrative embodiments of the invention. In FIG. 5 a, an analogmultiplier is configured as a squarer, with both its inputs derived fromthe output of the balanced photoreceiver in FIG. 4.

In FIG. 5 a, the frequency multiplied is an analog RF multiplier (e.g.,Analog Devices AD834 or AD835) configured as a frequency doubler. Thisconfiguration performs a squaring function on a sinusoidal input toproduce a sinusoid at twice the frequency. A bandpass filter eliminatesthe offset introduced by the squaring process. Another version of theembodiment of FIG. 5 a is shown in FIG. 5 b. In FIG. 5 b, the frequencydoubler splits the input sinusoidal waveform into two waveforms with arelative phase difference of 90 degrees.

In FIG. 5 b, a phase-shifting power splitter is used to generate a pairof sinusoidal signals with 90-degree phase difference, approximatelyindependent of frequency. The two outputs are fed into an analogmultiplier to produce a sinusoid at twice the frequency. Thephase-shifted sinusoids are multiplied together to produce a sinusoid attwice the frequency. Unlike the FIG. 5 a embodiment, the embodiment ofFIG. 5 b does not require a bandpass filter, because no offset isintroduced by the multiplication process.

FIG. 6 shows two additional frequency multiplier embodiments that aredesigned for clock frequency doubling according to illustrativeembodiments of the invention. In the embodiment in FIG. 6 a, azero-crossing detector first converts the sinusoidal output of thesample clock interferometer into a square wave. A delayed version of thesquare-wave is then exclusive-ORed with itself to produce an ADC clockwith twice the frequency of the input sinusoidal waveform. The delayedpulse train is generated by a digital delay line, set for a delay τequal to ¼ of the shortest interpulse interval.

In the embodiment in FIG. 6 b, a pair of sinusoidal signals with a90-degree phase difference is generated with a phase-shifting powersplitter. Specifically, the input sinusoidal waveform is split by apower splitter into two waveforms with a relative phase difference of 90degrees. These signals are then converted into square waves that areexclusive-ORed to produce the frequency-doubled ADC clock. Thisembodiment has the advantage that the sample clock maintains a constant50% duty cycle over a wide frequency range. To enhance most pipelinedanalog-to-digital converters performance, they are driven with a dutycycle close to 50%.

The delay required for frequency multiplication of the interferencesignals can be realized in the optical domain as well as the electricaldomain, as illustrated by the embodiments of the sample clock generatorsshown in FIGS. 7-9. These embodiments take advantage of the phaserelationships among optical signals that combine within interferometersbased on N×N fiber couplers.

For example, the phase-splitting interferometer in FIG. 7 is fabricatedby replacing the output 2×2 coupler of a conventional Mach-Zehnderinterferometer (having an optical path imbalance equal to ΔL) with a 3×3coupler. When the 3×3 coupler has a specific splitting ratio (˜about29.3%:˜about 41.4%:˜about 29.3%), the interference signals formed at twoof its outputs have a relative phase difference of 90 degrees. In theembodiment of FIG. 7, the power-splitting ratio about 29.3%:˜about41.4%:˜about 29.3% is chosen to provide two equal-amplitude outputs withquadrature phase. These two outputs are multiplied and passed through azero-crossing detector. Thus, the electrical signals can be processedseparately and mixed in an analog multiplier to form a frequency-doubledsinusoidal waveform. Alternatively, as shown in FIG. 8, thephase-shifted optical signals can be processed by using the digital XORtechnique (discussed above) to produce a frequency-doubled ADC sampleclock.

In systems in which balanced photodetection is required to reducedegradation of the clock signal caused by laser intensity noise, theembodiment in FIG. 9 may be preferred. As shown, two pairs ofphase-shifted optical signals with opposite polarities are formed byreplacing the output 2×2 coupler of a conventional Mach-Zehnderinterferometer with a 4×4 coupler that splits the optical power equallyamong its four outputs. This embodiment is based on a 4×4phase-splitting interferometer that provides a pair of balanced outputswith quadrature phase relationship. As in the FIG. 8 embodiment, theresultant optical signals are processed digitally using XOR techniquesto produce a frequency-doubled ADC sample clock.

FIG. 10 illustrates yet another embodiment of the sample clockgenerator. Unlike the embodiments in FIGS. 4-9, this embodiment producestwo separate ADC sample clocks with a quadrature phase relationship.These Sine and Cosine clocks can be used to acquire OCT interferencesignals from the main interferometer on parallel ADC channels at thefundamental sampling frequency set by the optical path imbalance (ΔL) ofthe sample clock interferometer.

Complex Fourier transformation of OCT signals permits reconstruction ofthe depth profile of the sample, while suppressing image artifacts thatarise from complex conjugate ambiguity. SS-OCT systems that reconstructdepth profile via Fourier transformation of real-valued interferencesignals suffer from artifacts generated by the superposition ofreflectors offset by equal distances on either side of the referencereflector. As shown in FIG. 11, an analogous optical phase-splittingmethod can be used to collect quadrature (complex) signals from the maininterferometer by using a pair of ADC converters clocked simultaneouslywith the same ADC clock.

In SS-OCT systems based on an FDML laser, precise control of both the acdrive waveform, which sets the laser repetition rate, and dc bias offrequency-tuning element, which sets the center wavelength of the sweep,is required to attain high signal-to-noise and wide dynamic range. Inone embodiment, the optimum ac drive frequency is defined as thefrequency at which the instantaneous linewidth of the laser is aminimum, which occurs when the round-trip time in the cavity and theperiod of the waveform match. At this frequency, when measured at thetime t=τ at which the laser scans through the zero-dispersion wavelengthof the optical delay element (typically 1310-1315 nm), the instantaneousRMS amplitude Φ(t) of the interference signal at the output of thesample clock interferometer's photoreceiver reaches a maximum.Therefore, the optimum drive frequency can be found by adjusting thedrive frequency to maximize Φ(τ).

FIG. 12 shows one of the preferred embodiments of a digital feedbackloop, which is based on a microcontroller that records Φ(t) with ananalog-to-digital converter at the time indicated by transmission of thepulse through a fiber-Bragg filter with a narrow passband (typically<1nm) at 1310 nm. The microcontroller adjusts the frequency of alow-jitter, frequency-agile DDS waveform generator until the recordedvalue of Φ(t) attains its maximum value. With respect to the embodimentof FIG. 12, the clock fringe control signal is obtained by detecting theinstantaneous RMS amplitude of the bandpass-filtered interference signalfrom the sample clock generator's photoreceiver. The RMS amplitude issampled by the control ADC at the time at which the frequency tuningelement scans through the zero-dispersion wavelength (1310 nm) of theoptical delay element in the FDML laser.

Turning now to FIG. 13, the figure illustrates how the instantaneous RMSamplitude of the sample clock interference signal varies at the optimumadjustment frequency and at frequencies above and below the optimum. Thefrequency of the waveform can be updated either continuously or atintermittent intervals determined by the maximum drift of the laser. Inaddition to its ac drive waveform, the dc bias of the frequency-tuningelement is adjusted to achieve optimum performance of the FDML-basedSS-OCT system.

One embodiment of a digital control loop for optimizing the dc bias isshown in FIG. 14. That is, the loop adjusts the amplitude of the DC biasuntil the time at which the frequency tuning element scans through thezero-dispersion wavelength (1310 nm) coincides with a fixed delay afterthe ac drive waveform crosses zero. This loop adjusts the dc bias suchthat the wavelength scan of the laser starts at a fixed wavelength,regardless of environmental influences that alter the voltagesensitivity of the tuning element. The same fiber Bragg filter as thatemployed in the frequency optimization control loop (FIG. 13) isemployed as a wavelength reference. By adjusting the dc bias via adigital-to-analog converter (DAC), the microcontroller maintains thetime interval at a constant level between the zero-crossings of the acdrive waveform from the DDS generator and the edge of the pulsegenerated by a comparator at the output of a photoamplifier connected tofiber Bragg filter.

The relationship and commercial feasibility associated with waveformgeneration, filter design, and laser behavior is important to considerwhen implementing the systems disclosed herein. Although (1) sinusoidalwaveforms are easy to generate with inexpensive DDS integrated circuitsand (2) most high-speed tunable filters with highly resonant responsesoperate best with sinusoidal actuation, this beneficial application ofsinusoids does not extend to all lasers. For example, lasers withlinear, rather than sinusoidal wavelength sweeps, provide higherperformance light sources for SS-OCT systems. With sinusoidal wavelengthsweeping, the instantaneous sampling clock frequency varies over a widefrequency range in proportion to the slope of the sine wave over itsperiod. Typically, precision high-speed analog-to-digital convertersaccept clock frequencies over a prescribed range (e.g., about 40-about210 MHz). Consequently, the effective duty cycle over whichinterferometric measurements can be acquired is, typically, limited toabout 33%. In addition, the Nyquist sampling frequency variescontinuously and rapidly in proportion to the sampling clock frequency.The use of tracking filters and the linearization approaches describedherein in various embodiments overcome this effective duty cycle limit.

Therefore, in one embodiment, to avoid aliasing, which results inobjectionable foldover artifacts in OCT images, the cut-off frequency ofthe anti-aliasing filter applied to the interference signal beforeanalog-to-digital conversion is configured to track ½ (or less) of theinstantaneous sampling frequency. Suitable tracking filters can beassembled by using, for example, varactor-tuned LC circuits. However,proper synchronization of the tracking controller requires complexdigital or analog control circuitry and to achieve the requiredsharpness, the filter is typically built from multiple stages withnarrow component tolerances. In contrast, linearizing the wavelengthsweep of the tunable filter over a large fraction of the wavelengthsweep can provide an alternate solution in some embodiments.

Using the Mach-Zehnder clocking methods described herein, ahigh-duty-cycle linear wavelength sweep produces a large number ofsample clock pulses with a narrower frequency distribution than asinusoidal wavelength sweep. Thus, higher speed imaging can be achievedwith less foldover artifacts at lower maximum data acquisition speeds.Unfortunately, linear actuation of commercially available Fabry-Perottunable filters at high speeds is difficult to achieve usingconventional triangular or ramp waveforms, because such broadbandwaveforms contain frequencies that excite strong resonant behavior ofthe actuators. Excitation of the filters with ramp or triangular drivewaveforms produces near-sinusoidal oscillations at the mechanicalresonance frequency rather than the desired linear scan.

As illustrated by the measured frequency response in FIG. 15,piezo-actuated filters typically exhibit the mechanical resonance with ahigh quality factor (Q=4-8) at frequencies in the 40-75 KHz range. Toachieve triangular or ramp excitation of these filters, the drivewaveform is tailored to provide linear mechanical response over anextended period while compensating for the highly non-uniform amplitudeand phase responses of a given filter.

Further, FIG. 16 illustrates a novel means of synthesizing a drivewaveform based on the summation of harmonically related sinusoidsaccording to the principal of Fourier synthesis. The period ofnear-linear amplitude decay of the drive waveform can be extendedsignificantly by forming the weighted sum of only 2 or 3 harmonics ofthe fundamental sine wave. The example waveforms are shown for f₀=45KHz. A first advantage of this method is that the fundamental andharmonic frequencies of the sine waves can be chosen to avoid the strongresonances in the mechanical response of the filter. In turn, a secondadvantage of this method, as illustrated in FIG. 17, is that only asmall number of harmonics are required to synthesize either smoothedtriangular or ramp waveforms. In addition, a third advantage is that theamplitudes and phases of the component sine waves can be tuned tocompensate for large non-uniformities in the amplitude and phaseresponses of tunable filter.

With respect to FIG. 17, the outputs of two phase-locked digital-directsynthesis (DDS) sine wave generators are summed and amplified to formthe drive waveform of the piezo-actuator. The phases and amplitudes ofthe DDS generators are adjusted to obtain the maximum duty cycle andlinearity of the portion of the drive waveform during which theinterferometric signals are sampled.

A specific embodiment of a tunable-filter linearizing circuit based ontwo phase-locked digital direct synthesis (DDS) sine-wave generators isshown in FIG. 18. The circuit is designed to generate smoothed rampdisplacement of a piezo-actuated Fabry-Perot filter with the frequencyresponse shown in FIG. 15. The primary excitation frequency of thefilter (about 45 KHz), which sets the repetition rate of the laser, istypically selected such that both this frequency and its second harmonic(about 90 KHz) are located outside of the major resonant peaks of thefilter response. In practice, the relative amplitudes of the about 45KHz and about 90 KHz sine waves are adjusted to obtain the narrowestrange of clock frequencies during the falling portion of the drivewaveform.

This tuning process can be performed in real time with an oscilloscopeset to display the gated Fourier transform of the clock signal. Testresults demonstrate that, compared to the conventional sinusoidal drivewaveform, the dual-sinusoidal harmonic drive waveform reduces themaximum clock frequency by about 30% and clock frequency span by afactor of 3, while maintaining the same about 100 nm sweep range. Theseimprovements increase the signal to noise ratio of the system and reducecertain artifacts.

It should be appreciated that various aspects of the claimed inventionare directed to subsets and substeps of the techniques disclosed herein.Further, the terms and expressions employed herein are used as terms ofdescription and not of limitation, and there is no intention, in the useof such terms and expressions, of excluding any equivalents of thefeatures shown and described or portions thereof, but it is recognizedthat various modifications are possible within the scope of theinvention claimed.

1-20. (canceled)
 21. An apparatus for tunable filter linearizationcomprising: a mechanically actuated tunable filter; and a pair ofphase-locked sine-wave generators having an output in electricalcommunication with the mechanically actuated tunable filter.
 22. Theapparatus of claim 21 further comprising a power amplifier electricallyconnected between the mechanically actuated tunable filter; and the pairof phase-locked sine-wave generators.
 23. The apparatus of claim 21wherein the mechanically actuated tunable filter is one of a piezo and amicro-electromechanically actuated Fabry-Perot filter.
 24. The apparatusof claim 21 wherein one of the pair of phase-locked sine wave generatorsoperates at twice the frequency of the other of the pair of phase-lockedsine wave generators.
 25. The apparatus of claim 21 wherein each of thepair of phase-locked sine wave generators is a digital direct synthesissine wave generator.
 26. A method for tunable filter linearizationcomprising the steps of: providing a mechanically actuated tunablefilter; and generating an output signal from a pair of phase-lockedsine-wave generators; and applying the generated output signal to themechanically actuated tunable filter, wherein one of the pair ofphase-locked sine-wave generators operates at twice the frequency of theother.
 27. The method of claim 26 wherein the mechanically actuatedtunable filter is a Fabry-Perot filter.
 28. The method of claim 26further comprising the step of amplifying the generated output signalprior to applying it to the mechanically actuated tunable filter.
 29. Amethod for tunable filter linearization comprising the steps of:choosing a fundamental frequency that avoids the mechanical resonancefrequencies of a tunable filter; summing two or three harmonicallyrelated sinusoids of the fundamental frequency; and tuning theamplitudes and phases of the summed sinusoids to compensate fornon-uniformities in amplitude and phase responses of the tunable filter.30. The method of claim 29 wherein the summing is performed by Fouriersynthesis.
 31. The method of claim 29 further comprising collectingoptical coherence data using an electromagnetic radiation source withinwhich the tunable filter is a component thereof.
 32. The method of claim29 further comprising generating light from a laser having a cavity, thetunable filter disposed within the cavity.
 33. The method of claim 31further comprising generating an image using the collected opticalcoherence tomography data.
 34. A linearized tunable filter comprising: amechanically actuated tunable filter having a resonance frequency; and alinearization module in electrical communication with the mechanicallyactuated tunable filter, the linearization module: generates afundamental frequency that avoids the mechanical resonance frequenciesof the tunable filter; sums two or three harmonically related sinusoidsof the fundamental frequency; and generates amplitudes and phases of thesummed sine waves to compensate for non-uniformities in the amplitudeand phase responses of tunable filter.
 35. The linearized tunable filterof claim 34 wherein the step of summing uses Fourier synthesis.
 36. Thelinearized tunable filter of claim 34 further comprising a gain elementin optical communication with the mechanically actuated tunable filter.37. The linearized tunable filter of claim 36 further comprising anoptical delay element in optical communication with the gain element.